This invention relates to electronic systems, and more particularly to digital filters, and even more particularly to filters used in communication systems.
In wireless communication systems such as those according to the GSM and other specifications organized by the Third Generation Partnership Project (3GPP), gaussian minimum shift keying (GMSK) modulation is used for impressing a data signal to be transmitted onto a radio frequency (RF) carrier signal. The modulated carrier signal is then transmitted by a communication device, such as a mobile telephone, computer, etc.
GMSK modulation can be applied directly to a frequency synthesizer that generates the RF carrier signal in the transmitter of the communication device. Direct synthesizer modulation can give better RF signal-to-noise performance than a traditional quadrature (in-phase/quadrature-phase, or I/O) modulator. Nevertheless, the frequency synthesizer is often a phase-locked loop (PLL), and PLLs usually have a modulation bandwidth (BW) that is too low for the GMSK modulation specified for the transmitter (TX). Even so, a pre-emphasis filter can effectively increase a PLL's modulation BW.
A typical single-point polar modulator for GMSK is built up as depicted in FIG. 1, which is a block diagram of a single-point GMSK polar modulator 100 that can be used in a communication device 10, such as a mobile telephone or other communication device. A pre-emphasis filter 110 is used to compensate for the frequency response characteristics of a phase-locked loop (PLL) 120, which includes a phase detector 121 that receives a reference clock signal of the carrier frequency, a loop filter 123, a voltage controlled oscillator (VCO) 125 that produces the modulated carrier signal, a divider 127, and a sigma-delta modulator 129. Symbol data to be impressed on the carrier signal is provided to a filter 130 having a frequency response suitable for GMSK modulation, such as a digital finite-impulse response (FIR) filter, and the spectrally shaped symbol data is provided to a suitable scaler 140. The pre-emphasis filter 110 can also be a digital filter.
FIG. 2 illustrates a frequency response (i.e., output signal magnitude with respect to frequency) of a pre-emphasis filter 110 and a frequency response of a PLL 120. As illustrated, the frequency response of the pre-emphasis filter is intended to compensate the frequency response of the PLL, which is to say that the combined frequency response is more or less flat (i.e., more or less constant magnitude with respect to frequency). In previous solutions, the filter coefficients of the digital pre-emphasis filter have been pre-calculated and stored in a memory, from which they have been fetched when needed.
For example, U.S. Patent Application Publication No. US 2004/0183602 describes measuring an actual value of at least one filter component in a PLL and a controller for determining at least one adaptive filter coefficient using the measured value to compensate for deviation of the measured value from an ideal value. The adaptive filter coefficients are obtained from look-up tables in order to reduce computational load, and generic methods of generating filter coefficients for a look-up table (or for direct use) are disclosed.
S. T. Lee et al., “A Quad-Band GSM-GPRS Transmitter with Digital Auto-Calibration”, IEEE J. Solid-State Circuits, vol. 39, no. 12, pp. 2200-2214, December 2004, describes a modulator with auto-calibration, in which data to be transmitted is sent through a modulation filter in parallel with a pre-distortion filter. The filter outputs are summed together and sent to a delta-sigma modulator, with the pre-distortion filter branch multiplied by a DC signal selected from a look-up table in an “auto-calibration process”.
In real applications, however, matching the frequency responses of the pre-emphasis filter and the PLL is always less than perfect. Mismatches and other variations often cause higher-order modulation phase errors toward the edges of the usable operation area, and FIG. 3A is a plot of an example of such behavior of phase error with respect to frequency. Such phase error behavior increases the risk of exceeding regulatory or other specification limits on the performance of a modulator.
Mismatch can arise from variations in the manufacturing process, temperature sensitivities, etc. of the modulator components. FIG. 3B depicts representative phase error behaviors of three different modulators, Units 1, 2, 3, that illustrate the effect of manufacturing process variations. As for temperature sensitivity, FIG. 3C depicts representative phase error behaviors of a modulator at three different temperatures, Cold, Normal, Hot, that illustrate the effect of temperature variations.
Mismatch can also arise because the PLL's VCO usually uses different divider ratios (i.e., changed numerical content of integer and/or fractional dividers) in order to enable the modulator to operate through a wide frequency band. The different divider ratios correspond to different frequency ranges, which may be called sub-bands, in the wide frequency band. FIG. 3D illustrates a representative saw-tooth-shaped behavior of the phase error over a wide frequency band that is observed with a PLL having a VCO that covers the wide band with a number of contiguous narrower sub-bands. The “tooth length”, which is to say, the distance in frequency between peaks of the phase error, typically corresponds to the length of a VCO sub-band and is caused by imperfect tunings of components that cause the loop response to change faster inside the sub-band than between sub-bands.
Multiple sets of pre-computed coefficients for the pre-emphasis filter 110 can be used to approximate an adjustable response of the filter. This is illustrated by FIG. 4, which depicts phase error with respect to frequency for three different sets of filter coefficients that yield respective minimal phase errors at respective different frequencies. The set of coefficients used for a given frequency would be the set that yielded a phase error less than the phase error of any other set at that frequency.
For example, U.S. Pat. No. 7,477,686 generally describes generating and selecting filters with filter parameters, e.g., for a phase modulator, using stored look-up tables for calculating the filter parameters or interpolating between two or more known sets of filter parameters. Interpolation is discussed only on a general level, and the patent focuses on measuring direct parameters of a PLL (such as gain, frequency response, etc.) and generating the filter based on that.
U.S. Pat. No. 7,912,145 describes an adaptive filter for a fractional-N sigma-delta modulator based on measuring frequency after the modulation frequency and comparing it to the signal after the loop filter. A suitable set of filter coefficients is calculated to adjust for variations in the analog parts of the PLL, although the patent does not explain how the calculations performed once it has determined what kind of response is needed from the pre-emphasis filter.
Nevertheless, such prior solutions require a vast amount of numerical data that has to be first pre-computed and then inspected manually for possibly needed numerical adjustments in coefficients and/or signal scaling. Thus, such solutions tend to be memory consuming and also error prone during the design phase of a modulator.